AN2623 Application note
Evaluation board for off-line forward converter based on L5991
Introduction
This application note gives a practical example of a 160 W, isolated, forward converter using the L5991, high frequency current mode PWM controller. Design procedures for both the power stage and controller are presented. Generally for this power level the norm ICE61000-3-2 imposes the use of a PFC preregulator stage, but some countries do not require compliance to this norm. The forward converter presented here does not have a PFC. Figure 1. 160 W off-line forward converter, evaluation board
October 2007
Rev 1
1/25
www.st.com
Contents
AN2623
Contents
1 2 3 Basis of forward topology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 Main characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 Design circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
3.1 3.2 3.3 3.4 3.5 Primar y controller: L5991 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 Output filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7 Output diodes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8 Power transformer design and MOSFET choice . . . . . . . . . . . . . . . . . . . . 9 Feedback loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
4
Experimental results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
4.1 4.2 4.3 4.4 4.5 4.6 High frequency ripple of output voltage and load regulation . . . . . . . . . . 16 Dynamic load test . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18 Star t-up behavior . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19 Wake-up time . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 Shor t circuit test . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21 Thermal measurement and global efficiency . . . . . . . . . . . . . . . . . . . . . . 22
5
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
2/25
AN2623
List of figures
List of figures
Figure 1. Figure 2. Figure 3. Figure 4. Figure 5. Figure 6. Figure 7. Figure 8. Figure 9. Figure 10. Figure 11. Figure 12. 160 W off-line forward converter, evaluation board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 Basic forward converter topology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 Reset circuits . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4 Electrical schematic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6 Vds and Ids of STW12NK90Z in full load condition at different input voltages . . . . . . . . . 15 High frequency ripple of output voltage in full load condition at different input voltages. . . 16 Output voltage behavior against the load and the Vin . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 Behavior of system under dynamic load at different input voltages . . . . . . . . . . . . . . . . . . 18 Behavior of system under dynamic load at different input voltages . . . . . . . . . . . . . . . . . . 19 Wake-up time of the system at different input voltages . . . . . . . . . . . . . . . . . . . . . . . . . . . 20 Behavior of the system in short circuit condition at different input voltages . . . . . . . . . . . . 21 Efficiency of the system . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
3/25
Basis of forward topology
AN2623
1
Basis of forward topology
A forward converter is typically used in off-line applications in the 100 W - 300 W power range. A simplified schematic of the forward converter can be seen in Figure 2. Figure 2. Basic forward converter topology
D1 + D2 Vd
Reset Circuit
L +
C
V0 _
_
A natural limitation of the forward converter is the need to completely reset the transformer, cycle by cycle, before the next MOSFET switches on. Different circuits are used for this purpose with advantages and drawbacks. The two simplest and most commonly used reset schemes are: the RCD reset circuit and the reset auxiliary winding both shown in Figure 3 (a-b). In the design presented in this document, the reset winding was used. It is advantageous with respect to efficiency because the energy stored in the magnetizing inductor goes back to the input and is not lost as using an RCD snubber net. The drawback of the reset circuit is that, generally, a higher voltage Power Mosfet is needed. In the present design a 900 V MOSFET was used. Figure 3. Reset circuits
CR
RR
N1
N2
NR
N1
N2
DR
DR
(a)
(b)
The primary controller IC used is the L5991. It is based on a standard current mode PWM controller and includes features such as programmable soft start, adjustable duty cycle limitation and a standby function that reduces the switching frequency when the converter is lightly loaded. The standby function, in this case, is not used to prevent the transformer from saturation. The output voltage regulation is obtained through a voltage reference and an error amplifier (TL1431) placed at the secondary side. A charge pump connected to an auxiliary winding guarantees a stable supply at the controller itself.
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AN2623
Main characteristics
2
Main characteristics
The design procedure is presented in this section and we will refer to the electrical schematic in Figure 4. The power supply electrical specifications are shown in Table 1 below. Table 1. Input and output parameters
Input parameters Vin fline Input voltage Line frequency Output parameters Vout Iout Pout Vout% Vout HF TA max Output voltage Output current Output power Efficiency at full load Max tolerance on output voltage Max output voltage ripple at switching frequency Maximum ambient temperature 35 V 4.5 A max continuous, 0.45 A min 160 W max 80% 3% 350 mV 70 C 88 ÷ 290 VRMS 50/60 Hz
5/25
3
4
3
1
CON2
NTC1
2
4
4
RA1 4.7 kOhm
U1
0
10 OHM1
Rg
1
3
1 2 3 Sync ST-BY RCT DC-LIM DC DIS Vref Vfb Vcomp PGND SS Vcc Vout Vc 10 9
C9 100 pF
16 15 14 13 12 11
Q1 STW12NK90Z-heatsink
2
1
R9 C10 100 pF 0.21 Ohm
7 8
2
6/25
LFILTERIN1
HT3545-472Y4R0-T01
Main characteristics
Figure 4.
FUSE1 DN1 D1 T1 BYT16P-400 heatsink L1 390 uH-5A
4A DIODE BRIDGE1 600V-6A +1 CB1
J1
1 2
47 nF X2 Cap
CA1 47 nF X2 Cap
25 16
C1 +
J2
0
STTH110 C2 100 uF, 450V + + R1 220 kOhm, 1/4W 100 uF, 450V C3
0
4 1 3 8 7 14
2 1
CON2
2.5 Ohm C4 330 uF, 450V +
270uF, ESR=42 mOhm, 50 V
Electrical schematic
C5 2.2 nF Y1 Cap
0
R2 220 kOhm,1/4W
0
TRAN_ISDN_06 R3 5.6 kOhm
D2 1N4148 33nF D3 15V C7 + 10uF, 20V
R4 C6 50 Ohm-1/2W
OPTO 1
R5 15 kOhm R6 1.2 kOhm ISO1 C8 6 nF R7 20 kOhm
CT1 4.7 nF
RDOWN1 3.6 kOhm
4 5 ISEN SGND 6
RUP1 4.7 kOhm
R8 2.2 kOhm
TL 431 . . ..1
3
R10 1.153 kOhm (+/- 1%)
L5991
0
C11 + C13 33 nF
C12 Title Size A Document Number Rev
1 nF
22uF, 25V
AN2623
0
0
0
0
0
0
0
AN2623
Design circuit
3
Design circuit
This section describes the design of the major parts of the circuit.
3.1
Primary controller: L5991
As previously stated, the L5991 is used as the primary controller and its components must first be selected. Refer to the L5991 datasheet for the choice of the two resistors (RA, RB ) and one capacitor (CT) which allows setting separately the operating frequency of the oscillator in normal operation (fosc) and in standby mode (fsb). In this application, it was established that the device must work at the unique frequency (in this case RB ) of 60 kHz in normal and in standby operation. This frequency is calculated using RA in the following formula: Equation 1
-f o s c = ------------------------------1-----------------------------C T ( 0.693 R A + K T )
where KT=160 and CT is calculated fixing the discharge oscillator capacitor time Td=5%Tsw Equation 2
T d = 30 10
9
+ K t C t C t = 4.7 n F, R A = 5.6 k
Establishing a Dmax = 50%, L5991 allows obtaining this last value in two different ways. The method that allows implementing the slope compensation, if needed, was used. The duty cycle limitation is obtained by applying the following voltage to pin3 : Equation 3
V3 = 5 2
( 2 Dma x )
V 3 = 2.17 V
fixing (refer to Figure 4) Rup=4.70 k, we can then immediately calculate Rdown=3.60 k.
3.2
Output filter
Admitting a max current ripple on the inductor ILout equal to 20% of IoutMAX, it is necessary to select an inductor value according to Equation 4: Equation 4
V2 m i n Vd i o d e Vo u t Dm a x L o u t = ------------------------------------------------------- ------------- L o u t= 342 H Io u t fs w
The RMS (root mean square) current through the inductor is given by Equation 5: Equation 5
IR M S L o u t = I
2 out
out + ------------- I R M S L o u t = 4.58 A 12
2
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Design circuit
AN2623
The peak current through the inductor is: Equation 6
I P e a k L o u t = I o u t + I L o u t = I P e a k L o u t = 5.4 A
According to these results, Lout was chosen as the Coil Craft's inductor PCV-1-394-05L whose inductance value is Lout=390 H. According to the max high frequency voltage ripple (VoutHF=350 mV) from the electrical specifications, the necessary minimum capacitor value (C1 in the Figure 4) and its maximum admitted ESR (Equivalent Series Resistance) are calculated as follows: Equation 7
1 Dm a x Vo u t 1 C o u t m i n = --------------------- -------------------- ---------------------- C o u t m i n = 4.5 F Lo u t Vo u t H F 8 f2 sw
Equation 8
Vo u t H F E S R m a x = --------------------- E S R m a x = 388 m Io u t
The RMS current through the output capacitor must not exceed the current rate of the selected capacitor and is calculated as: Equation 9
IR M S C o u t = I
2 RMS Lout
I
2
out
I R M S C o u t = 860 m A
According to these requirements a Cout=C1=270 F (capacitance value) 63 V (Voltage rate) ZL series Rubycon electrolytic capacitor was selected with an ESR of 42 m and max current capability of 1495 mA.
3.3
Output diodes
The maximum reverse voltages across the rectifier diode and the free wheeling diode (D1D2 in the Figure 2) can be calculated as: Equation 10
V1 m a x V d i o d e R = --------------- V d r o p F V d i o d e R = 328 V n
Equation 11
V1 m a x V d i o d e F = --------------- V d r o p R V d i o d e R n
VdropF and VdropR are, respectively, the voltage drop in the freewheeling diode and in the rectifier diode, when they are forward biased, and n=1.25 is the turn ratio between the primary and the secondary winding of the transformer. Considering that the voltage drops in the two diodes are the same, we can conclude from Equation 10 that VdiodeR = VdiodeF . The maximum RMS and the average currents through the rectifier diode are calculated as:
8/25
AN2623 Equation 12
IR M S d i o d e R = Io u t Dm a x
Design circuit
2 I 1 1 + ----- ---o-u-t I --- ---R M S d i o d e R = 3.2 A Io u t 12
I A V G d i o d e R = I o u t D m a x = 2.25 A
and for the free wheeling diode: Equation 13
Vd c m i n D m i n = D m a x ------------------ = 11.5 % I R M S d i o d e F = I o u t Vd c m a x ( 1 Dm i n )
2 1 1 + ----- -I-o-u-t = 4.23 A ------- 12 I o u t
Equation 14
I A V G d i o d e F = I o u t ( 1 D m i n ) = 3.825 A
In Equation 12, 13, and 14 the currents are calculated in Full Load condition considering the worst case for each diode and can be used to calculate the maximum power dissipation for each diode. To reduce the number of components, the size of the board, and to minimize power losses, the ST double fast recovery rectifier BYT16P-400 was selected. Although the two diodes inside the same package are always working complementarily, in order to choose the heat sink, the total power losses in the worst case can be calculated as if, instead of two diodes there is only one that flows through the whole current of the inductor: Equation 15
Pt o t D i o d e = Vt Io u t + Rd I
2 out
I o u t 2 1 1 + ----- ------------ = 5.53 W 12 I o u t
Considering TAmbMax = 70 C, the power losses just calculated, and the maximum junction temperature TJmax (see diode datasheets) of the selected diode, it is possible to determine the total thermal resistance of the diode: Equation 16
Tj m a x TA m b M a x R t h m a x = -------------------------------------------- = 15 C /W Pl o s s e s R
max
The Rthmax that results is lower than the max junction to ambient thermal resistance RthJ-A of the select diode, so a heat sink with thermal resistance RthSN 13 C must be added.
3.4
Power transformer design and MOSFET choice
Ideally in a forward converter, the energy flows forward from the primary side to the secondary side without any storage in the transformer. But the real transformer does not have infinite magnetizing inductance, so during the on-time of the power MOSFET some energy is stored in the magnetic core. The proper magnetic core and the primary winding turn number have to be selected in order to avoid core saturation. The proper magnetic core and primary winding turn number must be selected taking into account that some energy is also dissipated in the magnetic core.
9/25
Design circuit
AN2623
An empirical formula that gives an indication regarding the needed area Product for magnetic core that has to be selected for the application is shown in Equation 17: Equation 17
11.1 P i 4 A P m i n = -------------------------------------------- = 1.47 c m 0.141 B f s w
where B is maximum flux density swing in Tesla for normal operation and its typical value is within 0.2-0.3T in the case of the forward converter. This value has to be chosen in order to avoid saturation and to limit core losses; we chose 0.2T. The selected core is ETD39 (AP=2.2cm4, Ae=125 mm2) in N27 material. Considering this kind of core and the transformer's max temperature rise max=40 C, the maximum allowed total power loss is: Equation 18
T P L O S T t r a s f T O T A L = ---------------------- = 2.5 W Rt h C O R E
and the result of the relative max allowed core loss: Equation 19
2 PL O S T t r a s f T O T A L P f e = ------------------------------------------------------- = 1.21 W k1 + 2
(k1 is the B swing exponent relative at N27 material) so the real value of the maximum flux density swing is: Equation 20
P L O S T t r a s f T O T A L k 1 B = 2 --------------------------------k----------- = 0.146 T V k f 2s w e 0
1 ----
(k0 is the loss coefficient of N27 material, K2 is the frequency exponent of N27 material and Ve is the effective volume of ETD39's core). The minimum primary turns is given by: Equation 21
Vd c m i n Dm a x N 1 m i n = -------------------------------------- = 42 A e f s w B
and we chose N1=42. The turn ratio n between primary and secondary side is defined as: Equation 22
N1 Vd c m i n Dm a x Vd c m i n n = ------ = ----------------- = -------------------------------------- = 1.15 Vo u t + Vd r o p R N2 V2 m i n
where N1 and N2 are the number of turns of the primary and secondary side, Vout is the output voltage and VdropR is the diode rectifier voltage drop. The secondary turns number is N2=36. The magnetizing inductance of the primary side is given by: Equation 23
Lm = AL N
2 1
10
9
L m 3.8 m H
where AL is the inductance for turn square in nH/turns2.
10/25
AN2623 Considering that the total instantaneous current at the primary side is Equation 24
i1 t o t ( t ) = i 2 ( t ) + im ( t )
Design circuit
where i'2(t) is the secondary winding current during ton reported at primary side and im(t) is the magnetizing current. The magnetizing current expression is: Equation 25
Vi n m i n i m ( t ) = ---------------- t Lm
And its peak value is: Equation 26
Vi n m n f w -I m p k = -------------i-------------s--Lm DM A X
The peak value for the i`2(t) is: Equation 27
I
2pk
I 0 1 = I o u t + ------- - 2 n
And the value for i`2(t) at switch-on is: Equation 28
I
2min
I 0 1 = I o u t ------- - 2 n
The ripple current at the primary side is: Equation 29
I 1 = I m p k + I
2pk
I
2min
The rms value for the current at the primary side is: Equation 30
I1 t o t R M S =
2 1 2 D M A X ( I 2 m i n ) + I 1 I 2 m i n + -- I 1 3
In this case neglecting the im(t) , from (Equation 24), it is possible to write following formula: Equation 31
IR M S d i o d e R I 1 t o t R M S ----------------------------- = 2.75 A n
Considering (Equation 18) and (Equation 19), the maximum allowed copper power losses in the windings transformer can be immediately calculated. It is possible to select the diameter for primary and secondary winding; we have chosen d1=0.25 mm and d2=0.8 mm.
11/25
Design circuit
AN2623
Considering that for the L5991 the maximum allowed voltage value at the current sense (IS, pin n13) is 1 V, it is possible to determine the value of the current sense resistor (R9 in Figure 4): Equation 32
1 R 9 = --------------------- = 0.23 I1 t o t p e a k
where I1totpeak is the peak value of i1tot(t) . The maximum turn ratio k between primary and reset winding, as known in technical literature, in order to achieve the complete demagnetization of the transformer is the following: Equation 33
1 Dm a x k m a x = ---------------------- k m a x = 1 Dm a x
Choosing Equation 34
NR k = ------- = 0.96 N1
the necessary number of turns for reset winding is NR=41.The maximum reverse voltage and average current of reset diode are given by (IAVE-1magn is the magnetizing average current of the primary side): Equation 35
V R E V R = V D C m a x k + V D C m a x = 806 V
Equation 36
IA V E 1 m a g n I A V E R = ---------------------------------- = 0.11 A k
The Bipolar ultrafast diode STTH110 was chosen. Concerning the MOSFET the maximum drain voltage is: Equation 37
1 V d r a i n M a x = ( V d c M a x V d r o p R e s e t ) -- + V d c M a x = 838 V k
with VdropReset as the voltage drop in the reset diode. The max rms drain current is: Equation 38
Id r a i n R M S = It o t 1 R M S
so the Zener-protected SuperMESH Power Mosfet STW12NK90Z was chosen. Calculating the estimated total MOSFET power losses, it is easy to concludethat a substantial heatsink (around Rth5 C/W) is necessary.
12/25
AN2623
Design circuit
3.5
Feedback loop
Since current mode control is employed using the L5991 current mode controller, the power stage of the forward converter exhibits a single output pole due to the output capacitor and load combination, along with a zero due to the ESR of the output capacitor. The goal of the compensator is to achieve a slope of -20 db/decade for the closed loop gain, with a phase margin greater than 45 degrees at the crossover frequency. To achieve good dc regulation, a high low-frequency gain is another requirement for the compensator. For continuous conduction mode operation, the transfer function of the forward converter (power stage) is: Equation 39
s 1 + ---- z G 1 ( s ) = G 1 o -------------------s 1 + -----
p
where (referring to Figure 4)
-0 G1o is the power block gain and results in G 1 o = ---------------2 3 R9 V ou R 0 = -------------t R0 is the effective total load resistance of the controlled output defined as P0
n R
R9 is the current sense resistance n is the turn ratio between the primary and secondary side
1 z = ----------------------------------------E S Rc o u t Co u t 1 p = ------------------------R0 Co u t
In order to reach the objective previously stated at the beginning of this section, the feedback compensation network transfer function, using L5991, is obtained as: Equation 40
s 1 + ------z c 1 C ( s ) = C 0 ----------------- -1 + ---s-- s -p c
where (referring to Figure 4)
12- 10 -- C-- R --T C0 is the feedback block gain and results in C 0 = ----R----------------8--------R------C 3 5
3
CTR is the current transfer ratio of the optocoupler R5 is the upper resistance of the out voltage divider of feedback net R3 is the polarization resistance of the optocoupler C8 and R7 are the capacitance and the resistance of the TL431's feedback net
zc is the zero of the feedback net z c =
pc is the pole of the feedback net p c =
1 -------------------- to compensate p R7 C8
1 --------------------------------------- to compensate z 3 12 10 C 12
C12 is the capacitor connected at COMP pin of L5991
13/25
Design circuit Choosing the crossover frequency f c 0.1 f s w f c = 5 k H z it is possible to place
AN2623
compensator zero fzc around --c fz c = 1600 H z and the compensator pole fpc above 3 f c f p c = 15 k H z .
f 3
Considering R3=5.6 k R5=15 k CTR=1 were calculated, choose the following values: , , C12=1 nF, C8=6 nF, R7=20 k .
14/25
AN2623
Experimental results
4
Experimental results
The schematic of the tested board is given in Figure 4. The graphs in Figure 5 show the drain voltage and current at the minimum, nominal and maximum input mains voltage during nominal operation at full load. Figure 5. Vds and Ids of STW12NK90Z in full load condition at different input voltages
Vin=88 Vac
Vin=220 Vac
Vin=290 Vac
Pur ple: drain voltage Brown: drain current
15/25
Experimental results
AN2623
The drain peak voltage (570 V) assures a reliable operation of the STW12NK90Z with a good margin against the maximum BVDSS.
4.1
High frequency ripple of output voltage and load regulation
Figure 6 shows the high frequency ripple of output voltage at minimum, nominal and maximum input voltages. Figure 6. High frequency ripple of output voltage in full load condition at different input voltages
Vin=88 Vac
Vin=220 Vac
Vin=290 Vac
16/25
AN2623
Experimental results Apar t the voltage spike, the voltage ripple of the output (at full load) for every input voltage is given in Table 2. Table 2. Value of high frequency output ripple at full load condition
Vin(V) 88 220 290 VoutHF (mV) 176 240 324 VoutHF% 0.5 0.69 0.9
Figure 7 shows the behavior of the output voltage regulation against the load. It is easy to see from the graph that, changing the load, the output voltage is practically constant. Figure 7. Output voltage behavior against the load and the Vin
3 .7 48 3 .7 46 O tp t V lta e (V uu o g ) 3 .7 44 3 .7 42 3 .7 4 3 .6 48 3 .6 46 3 .6 44 3 .6 42 01 . 04 .5 Io t(A u) 23 . 45 . Vn 8 V i =8 Vn 2 0 i =2V Vn 2 0 i =9V
17/25
Experimental results
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4.2
Dynamic load test
The graphs in Figure 8 show the output voltage regulation against a dynamic load variation (between max load and 10% max load). Figure 8. Behavior of system under dynamic load at different input voltages
Vin=88 Vac
Vin=220 Vac
Vin=290 Vac
Blue: Vcomp (voltage on pin 6 of L5991) Green: output voltage Brown: output current
18/25
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Experimental results
4.3
Start-up behavior
Figure 9 shows rising slopes at full load of the output voltage at nominal, minimum and maximum input main voltages. As shown in the graphs, the rising times are fairly constant. Figure 9. Behavior of system under dynamic load at different input voltages
Vin=88 Vac
Vin=220 Vac
Vin=290 Vac
Blue: Vcomp (voltage on pin 6 of L5991) Green: output voltage Brown: output current
19/25
Experimental results
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4.4
Wake-up time
Figure 10 shows the waveforms with wake-up time measures at nominal, minimum and maximum input voltages. Obviously due to the circuit characteristics, the wake-up time is not constant but it is dependent on the input voltage. The measured time at 88 Vac, 220 Vac and 290 Vac are (respectively) 2.48 sec, 780 ms and 580 ms which are rather common values for this kind of power supply. Figure 10. Wake-up time of the system at different input voltages
Vin=88 Vac
Vin=220 Vac
Vin=290 Vac
Bl
20/25
V
AN2623
Experimental results
4.5
Short circuit test
All tests have been done at nominal, maximum and minimum input voltages. For all conditions the drain voltages are always below BVDSS. As clearly indicated in the waveforms, the circuit starts to work in hiccup mode. Because the working time and the idle time are imposed by the charging and discharging time of the auxiliary capacitor C11 (refer to Figure 4), they are proportional to the input mains voltage. Figure 11. Behavior of the system in short circuit condition at different input voltages
Vin=88 Vac
Vin=220 Vac
Vin=290 Vac
Blue: VCC Brown: output voltage Purple: drain voltage As expected the circuit protects itself as well.
21/25
Experimental results
AN2623
4.6
Thermal measurement and global efficiency
One of the most critical parts of the power supply is the MOSFET. As previously seen at the end of Section 3.4: Power transformer design and MOSFET choice a heatsink is necessary. To verify the correct thermal behavior of the MOSFET, it was checked at maximum load and maximum input voltage. The device reaches the thermal steady state of 94 C. Figure 12 shows the global efficiency in function of the input voltage for two values of load. From the graph, we can conclude that the board has good efficiency. In absolute terms the minimum value is around 80% for Vin=290 V and the maximum is around 86% for Vin=88 V. The technical requirements for this converter have been respected. Figure 12. Efficiency of the system
8 .0 70 8 .0 60 8 .0 50 8 .0 40 8 .0 30 8 .0 20 8 .0 10 8 .0 00 7 .0 90 7 .0 80 7 .0 70 7 .0 60 8 8 20 2 In u v ltag (V pt o e) 20 9
Go a E fic e c (% l bl f i ny )
L a =4 5 od . A L a =2 3 od . A
Table 3.
Item 1 2
Bill of material
Quantity 1 2 Reference 1 CB1 CA1 Properties TL 431 STMicroelectronics part 47 nF X2 Cap 47 nF X2 Cap 4.7 nF 270 F, ESR=42 m 50 V, electrolytic capacitor , 100 F, 450 V, electrolytic capacitor 100 F, 450 V, electrolytic capacitor 330 F, 450 V, electrolytic capacitor 2.2 nF, Y1 Cap 33 nF 10 F, 20 V electrolytic capacitor 6 nF 100 pF ceramic capacitor 100 pF ceramic capacitor 22 F, 25 V, electrolytic capacitor 1 nF ceramic capacitor
3 4 5
1 1 2
CT1 C1 C2 C3
6 7 8 9 10 11
1 1 1 1 1 2
C4 C5 C6 C7 C8 C9 C10
12 13
1 1
C11 C12
22/25
AN2623 Table 3.
Item 14 15 16 17 18 19 20 21 22
Experimental results Bill of material (continued)
Quantity 1 1 1 1 1 1 1 1 2 Reference C13 Diode bridge1 DN1 D1 D2 D3 FUSE1 ISO1 J1 J2 23 24 25 26 27 1 1 1 1 2 LFILTERIN1 L1 NTC1 Q1 RUP1 RA1 28 29 30 31 32 33 34 35 36 37 38 39 40 41 1 1 1 1 1 1 1 1 1 1 1 1 1 1 RDOWN1 R1 R2 R3 R4 R5 R6 R7 R8 R9 R10 T1 U1 Rg Properties 33 nF ceramic capacitor 600 V - 6 A BYT16P-400 STMicroelectronics part + heatsink STTH110 STMicroelectronics part Diode 1N4148 Diode Zener Vz=15 V 4A OPTO 1-PC817 CON2 CON2 Common Choke, 15 mA Inductor, 390 H-5A 2.5 STW12NK90Z - STMicroelectronics part + heatsink 4.7 k 4.7 k 3.6 k 220 k 1/4 W , 220 k 1/4 W , 5.6 k 50 -1/2 W 15 k 1.2 k 20 k 2.2 k 0.21 1.153 k Transformer L5991 STMicroelectronics part 10
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Revision history
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5
Revision history
Table 4.
Date 29-Oct-2007
Document revision history
Revision 1 Initial release Changes
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