AN1537 APPLICATION NOTE
A SIMPLE TRICK ENHANCES
L5991's STANDBY FUNCTION
by C. Adragna and G. Gattavari
This application notes describes a simple technique that allows improving the Standby function of the advanced PWM controller L5991. The price to pay for that is the addition of just two resistors and two diodes, but the benefit brought in terms of no-load consumption in mains-operated converters is worth this small fee. The effectiveness of the improved Standby function will be proved and assessed on a couple of existing designs.
Introduction L5991's Standby function is a valuable help in reducing light-load input consumption of offline converters and making them compliant with energy saving standards such as EnergyStar, Energy2000 and others. This function, optimized for flyback topology, is the ability of automatically - and abruptly - reducing the oscillator frequency (i.e. converter's switching frequency) as the converter's load falls below a defined threshold and restoring the normal oscillator frequency as the load increases and exceeds a second threshold. The frequency shift allows minimizing power losses related to switching frequency, which represent most of losses at light or no load, without giving up the advantages of a higher switching frequency at full load. Being the L5991 a current-mode controller [1], the output voltage (VCOMP) of its error amplifier (pin 6, COMP), except for an offset, is proportional to the peak primary current and then to the energy handled by the transformer cycle by cycle. It is then possible to deduce converter's load conditions by monitoring V COMP. Figure 1. L5991's Standby function operation: fsw vs. VCOMP locus (left) and VCOMP vs. Pin locus (right).
f sw Pin
fosc
Norma l operation
Unde rsho ot during transition
Normal operation
fosc
PNO P SB
Standby
Ove rshoo t d uring transition
f SB
fSB
1
Standby
2
VT 1
3
VCOMP
VT2
4
1
2
VT1
3
VCOMP
VT2
4
If the peak primary current decreases as a result of a decrease of the power demanded by the load and VCOMP falls below a fixed threshold (VT1), the oscillator frequency will be set at a lower value (fSB). If now the peak primary current increases and VCOMP exceeds a second threshold (VT2 > VT1) the oscillator frequency will be reset at the normal value (fosc). Since the frequency shift causes VCOMP to shift too but in the opposite direction for energy balance reasons, an appropriate hysteresis (VT2-VT1) is provided to prevent the oscillator frequency from switching back and forth between fSB and fosc. This operation is shown in fig. 1. The L5991 allows programming both the normal and the standby frequency. VT1 and VT2 are internally fixed but
May 2003 1/18
AN1537 APPLICATION NOTE
it is possible to adjust the thresholds in terms of input power level (PNO, PSB) by adding a DC offset on its current sense input (pin 13, ISEN). Reference [2] provides plenty of details on this function and its usage. There is a maximum abrupt frequency shift allowed, which is related to the amount of hysteresis: the theoretical maximum ratio of fosc to fSB is 5.59, however this value does not account for the dynamic changes of VCOMP during the transients resulting from the frequency shift. As a matter of fact, depending on the closed-loop characteristics of the voltage control loop and on the amplitude of the load change that causes the frequency shift, VCOMP may overshoot or undershoot before reaching its new steady-state value (see figure 1). If during a transient the other threshold is crossed, VCOMP may bounce from one threshold to the other and the switching frequency be unstable, going back and forth from one value to the other. As a result, the practical limit is less than the theoretical value, probably less than 4 and, at any rate, the control loop dynamics needs to be kept relatively slow to limit the aptitude of VCOMP to under- or overshooting. In [2] it is explained also that the addition of a DC offset on the current sense pin increases the maximum fosc to fSB ratio allowed. However, this technique is suitable for allowing a higher fosc with a given fSB. The lower limit on fSB is determined by other considerations: if it is in the audible range (< 16kHz), the transformer will very likely generate audible noise, especially at power levels where the frequency is about to shift back to fosc, because of the high peak current involved. Often, instead, for a given fosc an fSB as low as possible would be required to meet the latest design targets aimed at complying even with the most severe energy saving standards. In this case it would be desirable to have a very low frequency under no-load conditions, where the peak current is too small to be able to generate audible noise, and a frequency above the audible range at power levels where audible noise issues may arise. This is exactly the purpose of the modification to the oscillator proposed in the following section. Standby function improvement To realize the aforementioned function, the oscillator frequency needs to be dependent on converter's load conditions - the lower the load, the lower the frequency and vice versa - and only when this is useful, that is at light load. This can be done by adding few parts to the oscillator of the L5991, as shown in figure 2. Assuming a perfect matching of the two diodes (with a common-cathode dual diode like the BAV70 this is closer to reality), when VCOMP falls below 3V (oscillator's peak voltage) some of the current that charges CT is diverted to ground through RC, D1 an R'. In this way the rate of rise of the voltage across CT is slowed down and the oscillator frequency decreased, the lower VCOMP the lower the frequency. Instead, when VCOMP is greater than 3V D1 isolates RC and the oscillator frequency will be either fosc or fSB, like in the standard L5991 oscillator circuit. RA, RB and CT can be then calculated as usual with the formulae given in [1]; as to the determination of RC and R' please refer to the appendix. Figure 2. Oscillator modification to improve Standby function
additional parts
6 COMP 16 S_BY
D2 D1, D2 2 x 1N4148 or 1 x BAV70 R'
D1
L5991
4
Vref RA RB RC
2 RCT CT
D2 compensates for the temperature shift of the forward voltage drop VF of D1. Considering that the current flowing through the diodes is in the hundred A or less, D1 and D2 dissipate negligible power and only ambient temperature affects their VF. Assuming D1 and D2 match perfectly, neither oscillator frequency nor the point where RC comes into play will depend on ambient temperature. In real-world operation, considering also that D1 and D2 do not usually carry the same current, a minimum temperature effect can be observed.
2/18
AN1537 APPLICATION NOTE
The "frequency foldback" provided by the additional circuit starts in the neighborhood of VCOMP = 3V, that is a little before that the high-to-low frequency shift takes place. After the shift, VCOMP will be higher and then the switching frequency will be close or exactly equal to fSB, depending on the fosc to fSB ratio. In applications where the switching frequency needs not be tightly fixed for some specific reason there is no major drawback to this technique. The only point to take care of is that the oscillator frequency be in the audible range only when the peak current is so low that no sound may come from the transformer, even when it is made with normal construction techniques. This can be obtained simply by choosing fSB well above the audible range (e.g. fSB > 30kHz seems to be a good rule of thumb). The benefits, on the contrary, are considerable: 1) Very low switching frequencies are possible which, as already stated, will allow treating the power throughput as much efficiently as possible: MOSFET's capacitive losses, gate drive consumption and other parasitic losses will be minimized. See [2] for more details on them. 2) Since the additional components will be concerned with taking the oscillator frequency to very low values, the standby frequency fSB can be kept relatively high, thus reducing the abrupt frequency shift and eliminating the need for a slow feedback to prevent frequency instability. As already said, keeping fSB high has the positive side-effect of eliminating audible noise issues. 3) As a result of the faster dynamic response, start-up under no-load conditions is possible even with a minimum pre-load on the output. The dummy load represented by the feedback network, as well as bleeders, if used, can be minimized. The limit to the dummy load reduction is given by the collapse that the voltage delivered by the self-supply winding experiences with no load, which must not pull the supply voltage of the L5991 below the UVLO threshold. To evaluate how much this function modification improves converter's performance at light or no load, the 45W wide-range mains AC-DC adapter illustrated in [3] and the 80W power-factor-corrected AC-DC adapter described in [4] will be optimized following the guidelines revealed by the above considerations. The "European Code of Conduct on Efficiency of External Power Supplies", ECC in short, whose limits are summarized in table 1, will be assumed as the reference. Table 1. Limits envisaged by European Code of Conduct on Efficiency of External Power Supplies
Max. no-load Power Consumption Rated Input Power 0.3W and < 15W 15W and < 50W 50W and < 75W Phase 1 01.01.2001 1.0W 1.0W 1.0W Phase 2 01.01.2003 0.75W 0.75W 0.75W Phase 3 01.01.2005 0.30W 0.50W 0.75W
Optimization of a 45W, wide-range mains AC-DC adapter For reader's convenience, table 2 summarizes the electrical spec of the adapter under consideration. Please refer to [3] for a detailed description and full evaluation data. Table 2. 45W, wide-range mains AC-DC adapter: electrical specification of the original design
Input Voltage Range (Vin) Mains Frequency (fL) Maximum Output Power (Pout) Output Normal Operation Switching Frequency (fosc) Light Load Switching Frequency (fSB) Full-load Efficiency (@ Pout =45W, Vin = 88÷264Vac) Maximum no-load Input Power (Vin = 88÷264Vac) 88 to 264 Vac 50/60Hz 45W Vout = 18V 3% Iout = 2.5A max. Full load ripple 2% pk-pk 70kHz typ. 18kHz typ. > 80% < 1W
3/18
AN1537 APPLICATION NOTE
Since the full-load input power is greater than 50W, this adapter belongs to the third bracket envisaged by the ECC. Its no load consumption is 0.9W @264Vac and 0.7W @220Vac then it meets Phase 1 limit (1W) and is close to that of Phase 2 and 3 (0.75W) with almost no margin. Although the ECC specifies that the compliance test be done at the nominal voltage 230 Vac, in the pre-compliance test it is quite usual to refer to the consumption at 264 Vac, to account for production spread. With this criterion the adapter cannot be considered compliant with Phase 2 or 3 limits. The target of the optimization is then to make the adapter ECC-compliant in the above mentioned sense. Figure 3 shows the electrical schematic of the converter with the added and modified components highlighted. Only these changes will be discussed. Figure 3. 45W AC-DC adapter: electrical schematic of the modified circuit
F1 T2A250V NTC1 N.A. R 22 N.A.
BD1 DF04M C1 100 F 400 V
C17 N.A. D5 BYW29-200 C10 330 F 25 V C11 C15 330 F 220 nF 25 V
T1
18V/ 2.5A
88 to 264 Vac
R1 56 k
R3 2.2 M
D1 BZW06-154 N1 N2 D2 STTA106
C9 330 F 25 V
R2 56 k C3 100 nF
R6 330 k
R4 2.2 M
D3 1N4148
R7 1
C12 4.7 nF 2kV D4 1N4148
GND
R5 47 k
R8 22 k R9 27 k
R10 22
C2 47 F 25 V
N3
R23 N.A. VRE F ST-BY
4 3
D CC
14
DIS
8
VCC 9
VC OUT
10
R11 10
Q1 STP7NB60
16
R12 12 k R13 1 2 k
R17 4.3 k
R1 8 20 k
IC1
ISEN
13
R 14 1 k
L 5991
2
RCT C4 100 nF DC-LIM
D6 1N4148
15
12
5 VFB SS
7
6
11 COMP PGND
C8 100 pF
R1 5 0.47 1/2 W
IC2
1
P C905
2
R19 N.A.
C 14 8.2 nF
SGND
D7 1N4148 Rc 5.9 k
R16 100
7 4
C13 N.A.
R20 180 k
C5 3.3 nF
R' 8.2 k
C6 56 nF
C7 3.3 nF
6
3
R21 3.16 k
1)
The oscillator has been modified to maintain the same frequency under normal operation (70kHz) at full load and have a standby frequency equal to half the normal frequency (35kHz). The oscillator frequency with no-load will be 5kHz. Further details on the calculations can be found in the appendix. 2) The dummy load represented by the feedback components on the secondary side (170mW in the original design) has been reduced at 40mW: R21 has been increased from 348 to 3.16k and, consequently, R18 from 2.2 to 20k to maintain the same regulated output voltage. This reduces the current consumption of the divider from 7.2 to 0.8mA. Additionally, R19 which was to provide 1mA extra bias current to the reference of the PC905, has been taken out since it was not strictly necessary. 3) The frequency compensation of the voltage control loop (C7, C14, R20) has been modified so as to get a larger bandwidth - it has been almost doubled - and then a faster response. The main purpose of that is to allow a correct start-up of the converter even with no load, whereas a slow feedback (basically, a large C14) causes the system to try continuously to restart under these conditions. 4) R7 has been decreased from 4.7 to 1 , to prevent the supply voltage of the L5991 from going below the UVLO threshold with no load. To help this, the total consumption of the IC has been reduced by 0.3 mA by increasing R8 and R9 (from 5.6 to 22k and from 6.8 to 27k, respectively). Although with this change the voltage generated at full load is higher, it is still below the OVP threshold, set by R5 and R6, with a safe margin. These modifications are summarized in table 3.
4/18
AN1537 APPLICATION NOTE
Table 3. 45W, wide-range mains AC-DC adapter: list of modifications to the original design
Part R7 R8 R19 R 12 R13 R18 R19 Original value 4.7 5.6k 6.8k 24k 8.2k 2.2k 1.2k New Value 1 22k 27k 12k 12k 20k --Part R20 R21 RC R' C7 C14 D6, D7 Original value 5.6k 348 ----220pF 470nF --New Value 180k 3.16k 5.9k 8.2k 3.3nF 4.7nF 1N4148
45W AC-DC adapter: evaluation results The following diagrams compare the performance of the original design ("standard standby") with that of the modified one ("improved standby"). For reference, it has also been measured the input consumption after replacing the start-up circuit made up of R1, R2 and D3 with an active start-up circuit (see fig. 12). To be noted in figure 4, the no-load consumption is < 0.6W @ 264Vac, then the adapter under test meets the ECC limits, Phase 3 (< 0.75W @230Vac) with some margin even without the use of an active start-up circuit. Figure 4. 45W AC-DC adapter: light load input consumption comparison
Pin [W] 1.6 1.4 1.2 1 0.8 0.6 50
improved standby
Pout = 0.5 W fsw = 16 kHz Tamb= 25 C
Pin [W] 1.4
standard standby
Pout = 0.3 W fsw = 13 kHz Tamb= 25 C
standard standby
1.2 1 0.8 0.6 0.4 50
improved standby
improved standby w ith active start-up (*)
improved standby w ith active start-up (*)
100
15 0 Vin [Vac]
200
250
300
100
150 Vin [Vac]
200
25 0
300
(* ) refer to the circuit shown in the schematic of figure 12
(*) refer to the circuit shown in the schematic of figure 12
Pin [W] 1 0.8 0.6 0.4 0.2 0 50
improved standby
Pout = 0 W fsw = 5 kHz Ta mb= 25 C
st and ard standby
improved standby w ith active start-up (*)
100
150 Vin [Vac]
20 0
250
30 0
(* ) refer to the circuit shown in the schematic of figure 12
The diagram on the left in figure 5 shows the relationship between output current and switching frequency obtained with the modified oscillator. The oscillator frequency is not much affected by the input voltage, as shown also by the oscilloscope diagrams of figures 6 to 10: the internal propagation delay of the current sense pin is compensated by R3 and R4, then the changes of VCOMP (and, consequently, fsw) with the input voltage are negligible. The diagram on the right in figure 5 illustrates the effect of temperature on both the oscillator frequency and the no-load input consumption (@264Vac) in the temperature range 0-70C: the variation is very limited.
5/18
AN1537 APPLICATION NOTE
Figure 5. 45W AC-DC adapter: fsw vs. Iout (left); Pin (@ Pout = 0) and fsw vs. ambient temperature (right)
fsw [kHz] 100
Tamb = 25 C
fsw [kHz] 7
fsw Pin
Pin [W] 0.58
50 30 20 10
Vin = 110 Vac, 230 Vac
6
0.57
5
Pout = 0 W Vin = 264 Vac
0.56
4 5 3 0.001 0.01 0.1 Iout [A] 1 3 -20
0.55
0
20 40 Tamb[C]
60
0.54 80
Figure 6. 45W AC-DC adapter: waveforms @ Pout = 45W
RCT (pin 2 of L5991)
RCT (pin 2 of L5991)
Q1 Drain Q1 Drain
Vin = 110 Vac, Pout = 45W
Vin = 220 Vac, Pout = 45W
Figure 7. 45W AC-DC adapter: waveforms @ Pout = 7W, just after the abrupt frequency shift
RCT (pin 2 of L5991)
RCT (pin 2 of L5991)
Q1 Drain Q1 Drain
Vin = 110 Vac, Pout = 7W
Vin = 220 Vac, Pout = 7W
6/18
AN1537 APPLICATION NOTE
Figure 8. 45W AC-DC adapter: waveforms @ Pout = 0.5W
RCT (pin 2 of L5991)
RCT (pin 2 of L5991)
Q1 Drain Q1 Drain
Vin = 110 Vac, Pout = 0.5W
Vin = 220 Vac, Pout = 0.5W
Figure 9. 45W AC-DC adapter: waveforms @ Pout = 0.3W
RCT (pin 2 of L5991)
RCT (pin 2 of L5991)
Q1 Drain Q1 Drain
Vin = 110 Vac, Pout = 0.3W
Vin = 220 Vac, Pout = 0.3W
Figure 10. 45W AC-DC adapter: waveforms @ Pout = 0W
RCT (pin 2 of L5991)
RCT (pin 2 of L5991)
Q1 Drain Q1 Drain
Vin = 110 Vac, Pout = 0W
Vin = 220 Vac, Pout = 0W
7/18
AN1537 APPLICATION NOTE
Figure 11. 45W AC-DC adapter: load transient 0.1 2.5A @ 220Vac
Iout
Vout
COMP (pin 6 of L5991)
Optimization of a 80W AC-DC adapter with PFC Table 4 summarizes the items of the electrical spec of this adapter more relevant to this context. Please refer to [4] for full electrical spec and evaluation data, as well as for a detailed description. Table 4. 80W AC-DC adapter with PFC: electrical specification of the original design
Input Voltage Range (Vin) Mains Frequency (fL) Maximum Output Power (Poutmax) Output 90 to 265Vac 50/60Hz 80W Vout = 18Vdc 2% Iout = 0 to 4.5A Vripple 1% < 1% 65kHz 20kHz > 75% < 1W
Line and Load regulation Switching Frequency (Flyback, @ Pout = 80W) Switching Frequency (Flyback, @ Pout = 0W) Target Overall Efficiency (@ Pout = 80W, Vin = 90÷265Vac) Maximum No-load Input Power (Vin = 90 ÷265Vac)
The full-load input power is greater than 100W, then this adapter is actually out of the scope of the ECC. Its no load consumption (0.9W @265Vac, 0.7W @220 Vac), however, is within the Phase 1 limit (1W) and not far from that of Phase 2 and 3 (0.75W @230 Vac) for the highest power bracket. Again, the target of the optimization is to meet Phase 3 limit @Vin = 264 Vac. Figure 12 shows the electrical schematic with the added and modified components highlighted. Only these changes, which track those made in the previously considered design plus those related to the PFC stage, will be discussed
8/18
T1A D1 STTA106
F1 T4A T1B R2 68K D4 5 IC1 3 8 6 4 R4 33 Q1 STP9NB50 2 1 D5
LF1-A
LF2-A
D2
D3
R3A, B 1.5M + 1.5M C8 1F R7C 499K
C28 100F 450V
C30 33 nF 400V R7A,B 1.5M + 1.5M +
90 Vac to 264 Vac C1 0.22F 630V
C2 0.47F
ZR1 SI0300
C3 0.68F 630V
NTC1 10
LF1-B
LF2-B
L6561
7
C5 4.7nF C7 10nF
C4 4.7nF
R1 22K
C29 10F 25V R5A, B 0.56 // 0.56
R6A 22K
R21 A, B 10M + 10M R27 N.A. D9 BAV19 C23 N.A. D10 1N4148
Q3 STD1NB50
R22A, B 10K + 10K
R38 47
R20 100K T2C D12 A,B,C 3 x 1.5KE68 T2A R26 22 VCC VC OUT 8 IC2 13 15 7 SS SGND VFB COMP
OC1 4 1
TR4 BC547 D11 BYW51- 200 C18 100F 25V C17 47F 25V T2B C19 680F 25V C20 680F 25V C21 680F 25V
R28 1
L1 2.2 C6 100nF C27 C22 680F 25V C24 680F 25V ZR2 +18V / 4.5A
C12 100nF
R19 100K D6 Z18 D7 MUR1100 R23 22 R24 1K R25A,B 0.56 // 0.56 Q2 STP7NB80 R39 10M
R17 47K
R18 330K
GND_OUT C26 4.7nF
DI S 9 Isen DC-LIM PGND 10
14
R14 6.8K
R13 6.8K
DCC
3
L5991A
2 RCT 12 5 6 11
D8 1N4148
16
4
ST- BY
VREF
R15 15K R16 11K
C14 4.7F R37 27K
R32 4.3K
R11 27K
3
2 P C8 17A
R30 N.A.
TR3 BC557
TR1 BC547
R12 47K TR5 BC557
Figure 12. 80W AC-DC adapter with PFC: electrical schematic of the modified circuit
R8 150K C9 100nF C10 3.3nF C11 10nF
D13 1N4148
RC 7.5K
D14 1N4148
R33 33K C16 N.A.
C15 220pF
VR1 TL431C
TR2 BC547
C25 3. 3n
R31 20K
AN1537 APPLICATION NOTE
R9 150K
R10 100K
C13 3.3nF
R36 5.6K
R29 3.16K
9/18
AN1537 APPLICATION NOTE
1 The oscillator of the L5991 has been modified to maintain the same frequency under normal operation (65kHz) at full load and have a standby frequency of 36kHz. Also in this case the oscillator frequency with no-load is chosen 5kHz. Further details on the calculations can be found in the appendix. 2 Again, the dummy load represented by the feedback components on the secondary side has been reduced at about 40mW (see the same point of the previous design). 3 The frequency compensation of the voltage control loop (C7, C14, R20) has been modified so as to double the bandwidth and get a faster response (especially useful at start-up). 4 R28 has been decreased from 3.3 to 1, to prevent the supply voltage of the L5991A from going below the UVLO threshold with no load, while not exceeding the rating of both the L5991A and the L6561. 5 The multiplier bias resistors (R3A = R3B = 680k and R1 = 10 k) of the L6561 consume 103mW @264 Vac, then their value has been more than doubled (R3A = R3B = 1.5M and R1 = 22k). In this way their maximum consumption is reduced at 46mW @264 Vac. 6 The output divider of the PFC pre-regulator (R7A = R7B = 499k and R1 = 6.34k) gives origin to a consumption of 140mW @264 Vac. With the aim of reducing the consumption at 40mW maximum, while not degrading the dynamic OVP function, the feedback network has been modified as shown in figure 13. The capacitor bypasses the two 1.5M resistors during output voltage overshoots, resulting in a only slightly higher OVP threshold, as compared to the original one. The positive sideeffect is an improvement of the transient response of the PFC pre-regulator. The components for the frequency compensation of the error amplifier of the L6561 have been changed consequently. The modifications are summarized in table 5. Figure 13. Modified PFC pre-regulator's feedback and frequency compensation network
PFC out
R7A 1.5M R7B 1.5M
C30 33 nF 400V
C8 1F
R7C 499K
2
R6A 22K
L6561
Table 5. 80W AC-DC adapter with PFC: list of modifications to the original design
Part R1 R3A, R3B R 6A R7A, R7B R7C R 15 R 16 R 28 R 29 R30 Original value 10k 680k 6.34k 499k --10k 22k 3.3 348 1.2k New Value 22k 1.5M 22k 1.5M 499k 15k 11k 1 3.16k --Part R31 R33 R34 R35 R36 RC C13 C25 C30 D13, D14 Original value 2.2k 13k 510k 2.2k 10k --15nF 330nF ----New Value 20k 33k --shor ted 5.6k 7.5k 3.3nF 3.3nF 33nF / 400V 1N4148
10/18
AN1537 APPLICATION NOTE
80W AC-DC adapter with PFC: evaluation results The following diagrams compare the performance of the original design ("standard standby") with that of the modified one ("improved standby"). To be noted in figure 14, the no-load consumption is slightly less than 0.5W @ 264Vac, then the modified 80W adapter now meets the ECC limits, Phase 3 for the 15-50W bracket (< 0.5W @230Vac). Figure 14. 80W AC-DC adapter with PFC: light load input consumption comparison
Pin [W] 1,5 1,4 1,3 1,2 1,1 1 0,9 0,8 50 100 150 Vin [Vac] Pin [W] 1
standard standby
Pout = 0 W Tamb= 25 C
Pin [W] 1,4
standard standby standard standby
Pout = 0.5 W Tamb= 25 C
1,2
improved standby
Pout = 0.3 W Tam b= 25 C
1 0,8 0,6 0,4 50
improved standby
200
250
300
100
150 Vin [Vac]
200
250
300
0,8
0,6
improv ed standby
0,4
0,2 50
100
150 Vin [Vac]
200
250
300
Figure 15. 80W AC-DC adapter with PFC: fsw vs. Iout (left); fsw vs. Vin (right)
fsw [kHz] 100 50 30 20
Vin = 110 Vac
Tamb = 25 C
fsw [kHz] 18 16 14 12 10 8 6
Pout = 0.3W Pout = 0.5W
Tamb= 25 C
10 5 3 0.001
Vin = 230 Vac
4
Pout = 0W
0.01
0.1 Iout [A]
1
5
2 50
100
150 200 Vin [Vac]
250
300
11/18
AN1537 APPLICATION NOTE
Figure 16. 80W AC-DC adapter with PFC: Pin (@ Pout = 0) and fsw vs. ambient temperature
fsw [kHz] 12
fsw Pin
Pin [W] 0.34 0.32 0.3 0.28 0.26 0.24 80
fsw [kHz] 12
fsw Pin
Pin [W] 0.46 0.44
Pout = 0 W Vin = 230 Vac
10 8 6 4 2 -20
10 8 6 4 2 -20
Po ut = 0 W Vin = 110 Vac
0.42 0.4 0.38 0.36 80
0
20 40 Tamb[C]
60
0
20 40 Tamb[C]
60
The diagrams in figure 15 illustrate the relationship between output current, input voltage and switching frequency obtained with the modified oscillator. Unlike the previously considered design, in this case the internal propagation delay of the current sense pin is not compensated, then VCOMP changes slightly with the input voltage and then there is a variation of the oscillator frequency. The diagrams of figure 16 illustrate the very limited temperature effect on both the oscillator frequency and the no-load input consumption at both nominal voltages in the temperature range 0-70 C. Figure 17. 80W AC-DC adapter with PFC, flyback section: waveforms @ Pout = 80W
RCT (pin 2 of L5991)
Q2 Drain
Figure 18. 80W AC-DC adapter with PFC, flyback section: waveforms @ Pout = 30W
RCT (pin 2 of L5991) RCT (pin 2 of L5991)
Q1 Drain Q 1 Drain
Vin = 110 Vac, Pout = 30W
Vin = 230 Vac, Pout = 30W
12/18
AN1537 APPLICATION NOTE
Figure 19. 80W AC-DC adapter with PFC, flyback section: waveforms @ Pout = 0.5W
RCT (pin 2 of L5991)
RCT (pin 2 of L5991)
Q2 Drain
Q2 Drain
Vin = 110 Vac, Pout = 0.5W
Vin = 230 Vac, Pout = 0.5W
Figure 20. 80W AC-DC adapter with PFC, flyback section: waveforms @ Pout = 0.3W
RCT (pin 2 of L5991)
RCT (pin 2 of L5991)
Q2 Drain Q2 Drain
Vin = 110 Vac, Pout = 0.3W
Vin = 230 Vac, Pout = 0.3W
Figure 21. 80W AC-DC adapter with PFC, flyback section: waveforms @ Pout = 0W
RCT (pin 2 of L5991)
RCT (pin 2 of L5991)
Q2 Drain
Q2 Drain
Vin = 110 Vac, Pout = 0W
Vin = 230 Vac, Pout = 0W
13/18
AN1537 APPLICATION NOTE
Figure 22. 80W AC-DC adapter with PFC, flyback section: load transient 0.1 4.5A @ 220Vac
Iout
Vout COMP (pin 6 of L5991)
Conclusions A simple modification of the oscillator allows optimizing L5991-based converters to achieve a considerable decrease of the no-load consumption. This has been proved in a 45W AC-DC adapter and in an 80W power-factorcorrected adapter, where the optimization here proposed has brought a reduction of the no-load input consumption up to 400mW at high line. This is the result of concurrent improvements. The modification to the oscillator allows very low switching frequency under no-load conditions without a large fosc to fSB ratio. The resulting lower VCOMP jump provides more headroom for overshoots and undershoots, so that it has been possible to enlarge the bandwidth of the control loop without any frequency instability. In turn, the faster response due to the larger bandwidth has allowed the reduction of the dummy load on the output, without compromising correct start-up when the adapter is unloaded. The lower residual load and the loss reduction due to the very low oscillator frequency, results in a dramatic decrease of the power absorbed from the mains at no load. REFERENCES [1] "L5991/L5991A Primary Controller with Standby" Datasheet [2] "Minimize Power Losses of Lightly Loaded Flyback Converters with the L5991 PWM Controller" (AN1049) [3] "45W AC-DC Adapter with Standby Function" (AN1134) [4] "80W Power-factor-corrected AC-DC Adapter with Standby Using the L6561 and the L5991A" (AN1440)
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AN1537 APPLICATION NOTE
APPENDIX How to calculate RC and R'. As previously said in the section "Standby function improvement", there is no change for the calculation of the timing components CT, RA, RB because RC is disconnected by the series diode D1 when VCOMP > 3V. Then only the calculation of RC to get a given fmin, as well as that of R' needs to be taken into consideration. To this end it is necessary to consider first the timing equations that govern the operation of the oscillator. Refer to figure A1 for the equivalent circuit and the relevant waveform, which assume that D1 and D2 match. Figure A1. Equivalent circuit for oscillator calculations (left) and theoretical V(RCT) waveform (right)
1 f SB
6 COMP SW
3
L5991
4
Vref RA RC
2.5
VCOMP
V(RCT) 2
2 RCT CT
1.5
SW open SW closed
SW
{
open if V(RCT)> VCOMP closed if V(RCT)> VCOMP
= RA · CT ' = (RA // RC) · CT
1 0
T1
t
Tsw
As long as the oscillator voltage V(RCT) is less then VCOMP, D1 is reverse-biased (SW is open) and only RA contributes to the time constant ; moreover, the asymptote of the exponential curve is the reference voltage VREF (5V). When V(RCT) equals and exceeds VCOMP, D1 is forward-biased (SW is closed), then RC goes in parallel to RA to determine the time constant ' and the asymptote will be changed to: RC RA V R E F --------------------- + V C O M P --------------------- < V R E F R A + RC RA + RC Note that the new time constant ' is shorter, then it will be the lower asymptote the responsible for getting a longer oscillator period. The complete equation describing the oscillator is then VV + ( VR E F VV ) 1 e
t
V(RCT) =
where:
------------RC RC RA ' t > T1 V R E F --------------------- + V C O M P --------------------- --------------------- ( V R E F V C O M P ) e R A + RC R A + RC R A + RC
VR E VV 4 -T 1 = ln ------------------F------------------- = ln ---------------------------5 VC O M P VR E F V C O M P is the time when V(RCT) = VCOMP and RC comes into play (i.e. SW is closed), VV is the oscillator valley voltage (1V) and the time constants and ' are defined as shown in figure A1. To find a simple formula to calculate RC some simplifications are necessary. Usually T1 << Tsw = 1/fmin, then T1 will be neglected and it is possible to write:
---------RC RC RA ' V p k = V R E F --------------------- + V C O M P --------------------- --------------------- ( V R E F V C O M P ) e , 0R + R 0 RA + RC A C RA + R C TSW
where Vpk is the oscillator peak voltage (3V) and VCOMP0 the value of VCOMP under no-load conditions. But, if T1 << Tsw it will be also ' << Tsw and the exponential term in (A2) can be neglected. Then (A2) becomes:
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t T1
t T1
(A1)
(A2)
AN1537 APPLICATION NOTE
RC RA V p k = V R E F --------------------- + V C O M P --------------------0R + R RA + R C A C that, solved for RC, yields: Vp k V C O M P 3 VC O M P 0 R C = R A ------------------------------------0 = R A ------------------------------ . V R E F Vp k 2
(A3)
The maximum value for R' can be calculated by imposing that D2 be always forward-biased, that is its current never fall to zero, with VCOMP = VCOMP0. The current flowing through D2 is: VC O M P VF 0 I D 2 = ---------------------------------- I D 1 , R' while the maximum current flowing through D1 is: V p k VF ( V C O M P VF ) Vp k VC O M P 3 VC O M P 0 0 0 I D 1 x = -------------------------------------------------------------------- = ------------------------------------- = ------------------------------ . RC RC RC Then, imposing ID2 > 0 with ID1 = ID1x and solving for R', it is possible to obtain: VC O M P V F V C O M P VF 0 0 R' < R C ------------------------------------- = R C ---------------------------------- . (A4) Vp k VC O M P 3 VC O M P
0 0
Since ID1 and ID2 are typically in the hundred A, with good approximation it is possible to assume VF = 0.5V @ 25C in (A4). Worst-case scenario is at minimum operating temperature: -2.5 mV/C drift can be considered to take this into account. The point is now to calculate the value of VCOMP0. In [2] it is shown that in L5991-based flyback converters the following relationships holds true: V C O M P = 1.4 + 3 R s 2 P i n Vi n -------------- ------- T d e l a y + V o , fs w L p L p
(A5)
where Rs is the sense resistor (R15 in figure 3, R25 in figure 12), Pin the input power to the transformer, Lp its primary inductance, Vin the DC input voltage to the converter, Tdelay the internal propagation delay of the current sense pin (200 ns typ.) and Vo a DC voltage offset that can be applied to the current sense pin. In case the offset Vo is chosen equal to Vi n R s ------- T d e l a y , Lp the effect of the propagation delay is compensated and VCOMP will no longer depend on the input voltage Vin. That is the case of the 45W adapter, where the job is done by R3 and R4 along with R14. For the 80W adapter there is no compensation (Vo = 0) then there will be a slight dependence of VCOMP (and then of the oscillator frequency when RC is active) on Vin. This is confirmed by the experimental results. Figure A2. VCOMP voltage under no-load conditions: 45W adapter (left), 80W adapter (right)
45W Adapter
80W Adapter
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AN1537 APPLICATION NOTE
To calculate VCOMP0, Pin needs to be estimated considering the power processed by the transformer under noload conditions. It is possible to use the following formula: Pin = 1.25(Vout Ioutres + Vaux Iaux) , (A6)
where Ioutres is the residual output current (through the output divider, through the optocoupler's LED, as well as any residual dummy load resistor), Vaux is the voltage generated by the auxiliary winding that supplies the L5991 (and the L6561 in the 80W adapter) and Iaux the total consumption on this winding (quiescent current of the L5991, of the L6561 too in the 80W adapter, optocoupler's transistor current and the consumption of any circuit supplied by the reference voltage of the L5991 or directly connected to its supply bus). The coefficient 1.25 stems from an estimated efficiency of 80% that experiments have shown many flyback transformers feature under no-load conditions, almost regardless of their size, parameters and construction. It is obvious that the estimate of Pin is the weak point of this design procedure. Since it is essentially the asymptote of the oscillator waveform that governs the oscillator frequency, even some ten mV variation of VCOMP0 may change the frequency considerably. A small error in the assessment of VCOMP0, then, could lead to a value for RC that gives an fmin away from the target. This is why it is recommended to use the result of the calculation as a starting point that has to be checked on the bench and, in case, corrected after the experiments. For completeness, it must be said also that under no-load conditions VCOMP is far from being a perfect DC voltage, as shown in the diagrams of figure A2. 45W adapter example calculation With CT = 3.3nF, RA and RB are 12k each. The residual output load is 40mW, the total consumption of the auxiliary winding is estimated around Iaux = 10mA, that is 110mW assuming Vaux = 11V. Then, using (A6): Pin 1.25 (0.04 + 0.11) = 0.188W . VCOMP0 will be given by (A5), where it is assumed a perfect propagation delay compensation: 2 0. 188 V C O M P = 1.4 + 3 0.47 ----------------------------------------------- = 2.011 V . 3 6 0 5 10 4 00 10 From (A3):
3 3 2. 011 3 R C = 12 10 ----------------------- = 5.934 10 use standard value 5.9k. 2
From (A4), considering Tamb = 0C, then VF = 0.563V: 3 2.011 0.5 63 3 R' < 5.9 10 ----------------------------------- = 8.638 10 use standard value 8.2k. 3 2. 011 80W adapter example calculation With CT = 3.3nF, we get RA =15k and RB = 11k. The residual output load is 40mW, the total estimated consumption of the auxiliary winding is Iaux = 12mA, that is 120mW assuming Vaux = 10V. Then, using (A6): Pin 1.25 (0.04 + 012) = 0.2W VCOMP0 will be given by (A5): V C O M P = 1. 4 + 3 0. 28
0
2 - ---.2 -----------------------0-------------------- ------------------0------------------- = 1.616 V . - 375 2 --0 - 10 3 6 6 5 1 0 4 30 10 430 10
9
From (A3):
3 3 1. 616 3 R C = 11 10 ----------------------- = 7.612 10 2 From (A4) ), considering Tamb = 0 C, then VF = 0.563 V:
3 1. 616 0. 563 3 R' < 7.5 10 ----------------------------------- = 5.706 10 use standard value 5.6k. 3 1.616
use standard value 7.5k
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AN1537 APPLICATION NOTE
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