TSH330
1.1 GHz Low-Noise Operational Amplifier
Bandwidth: 1.1GHz (Gain=+2) Quiescent current: 16.6 mA Slew rate: 1800V/s Input noise: 1.3nV/Hz Distortion: SFDR = -78dBc (10MHz, 2Vp-p) Output stage optimized for driving 100 loads Tested on 5V power supply
Pin Connections (top view)
D SO-8 (Plastic Micropackage)
Description
The TSH330 is a current feedback operational amplifier using a very high-speed complementary technology to provide a large bandwidth of 1.1GHz in gain of 2 while drawing only 16.6mA of quiescent current. In addition, the TSH330 offers 0.1dB gain flatness up to 160MHz with a gain of 2. W ith a slew rate of 1800V/s and an output stage optimized for driving a standard 100 load, this device is highly suitable for applications where speed and low-distortion are the main requirements. The TSH330 is a single operator available in the SO8 plastic package, saving board space as well as providing excellent thermal and dynamic perfor mances.
NC 1 -IN 2 +IN 3 -VCC 4 SO8 _ +
8 NC 7 +VCC 6 Output 5 NC
Applications
Communication & video test equipment Medical instrumentation ADC drivers
Order Codes
Part Number TSH330ID TSH330IDT Temperature Range -40C to +85C Package SO8 SO8 Conditioning Tube Tape&Reel Marking TSH330I TSH330I
June 2005
Revision 3
1/19
TSH330
Absolute Maximum Ratings
1 Absolute Maximum Ratings
Table 1. Key parameters and their absolute maximum ratings
Symbol VCC Vid Vin Toper Tstg Tj Rthja Rthjc Pmax Supply Voltage
1
Parameter
Value 6
Unit V V V C C C C/W C/W mW kV kV V V kV kV mA
Differential Input
Voltage2
+/-0.5 +/-2.5 -40 to + 85 -65 to +150 150 60 28 830 2 0.6 200 80 1.5 1 200
Input Voltage Operating Free Air Temperature Range Storage Temperature Maximum Junction Temperature SO8 Thermal Resistance Junction to Ambient SO8 Thermal Resistance Junction to Case SO8 Maximum Power Dissipation (@Ta=25C) for Tj=150C HBM: Human Body Model (pins 1, 4, 5, 6, 7 and 8) HBM: Human Body Model (pins 2 and 3)
5 4
Range3
ESD
MM: Machine Model 6 (pins 1, 4, 5, 6, 7 and 8) MM: Machine Model (pins 2 and 3) CDM: Charged Device Model (pins 1, 4, 5, 6, 7 and 8) CDM: Charged Device Model (pins 2 and 3) Latch-up Immunity
1) All voltages values are measured with respect to the ground pin. 2) Differential voltage are non-inverting input terminal with respect to the inverting input terminal. 3) T he magnitude of input and output voltage must never exceed VCC +0.3V. 4) Short-circuits can cause excessive heating. Destructive dissipation can result from short circuit on amplifiers. 5) Human body model, 100pF discharged through a 1.5k resistor into pMin of device. 6) T his is a minimum Value. Machine model ESD, a 200pF cap is charged to the specified voltage, then discharged directly into the IC with no external series resistor (internal resistor < 5), into pin to pin of device.
Table 2. Operating conditions
Symbol VCC Vicm Parameter Supply Voltage 1 Common Mode Input Voltage Value 4.5 to 5.5 -Vcc+1.5V, +Vcc-1.5V Unit V V
1) T ested in full production at 5V (2.5V) supply voltage.
2/19
Electrical Characteristics
T SH 330
2 Electrical Characteristics
Table 3. Electrical characteristics for VCC= 2.5Volts, Tamb=+25C (unless otherwise specified)
Symbol Parameter Test Condition Min. Typ. Max. Unit
DC performance
V io Vio Iib+ IibCMR SVR PSR ICC Input Offset Voltage Offset Voltage between both inputs Vio drift vs. Temperature Ta m b Tmin. < Tamb < Tmax. Tmin. < Tamb < Tmax. -3.1 0.18 0.8 1.6 26 21 7 13 50 63 54 54 74 67 56 52 16.6 16.6 20.2 22 55 +3.1 mV V/C A A dB dB dB mA mA k k
Non Inverting Input Bias Current Ta m b DC current necessary to bias the input + Tmin. < Tamb < Tmax. Inverting Input Bias Current Ta m b DC current necessary to bias the input Tmin. < Tamb < Tmax. Common Mode Rejection Ratio 20 log (Vic/Vio) Supply Voltage Rejection Ratio 20 log (Vcc/Vout) Power Supply Rejection Ratio 20 log (Vcc/Vout) Supply Current DC consumption with no input signal Vic = 1V Tmin. < Tamb < Tmax. Vcc= 3.5V to 5V Tmin. < Tamb < Tmax. Vcc=200mVp-p@1kHz Tmin. < Tamb < Tmax. No load Tmin. < Tamb < Tmax. Vout= 1V, RL = 100 Tmin. < Tamb < Tmax. 104
Dynamic performance and output characteristics
Transimpedance Output Voltage/Input Current Gain in open loop of a CFA. For a VFA, the analog of this feature is the Open Loop Gain (AVD) -3dB Bandwidth Frequency where the gain is 3dB below the DC gain AV Note: Gain Bandwidth Product criterion is not applicable for Current-FeedbackAmplifiers 153 152 Vout=20mVp-p, RL = 100 AV = +1 AV = +2 AV = -4 AV = -4, Tmin. < Tamb < Tmax.
R OL
Bw
550
1500 1100 630 600 160
MH z
Gain Flatness @ 0.1dB Small Signal Vout=20mVp-p Band of frequency where the gain varia- AV = +2, RL = 100 tion does not exceed 0.1dB SR Slew Rate Maximum output speed of sweep in large signal High Level Output Voltage Low Level Output Voltage Vout = 2Vp-p, AV = +2, RL = 100 RL = 100 Tmin. < Tamb < Tmax. RL = 100 Tmin. < Tamb < Tmax. 1.5
1800 1.64 1.54 -1.55 -1.5 -1.5
V/s V V
V OH V OL
3/19
TSH330
Electrical Characteristics
Table 3. Electrical characteristics for VCC= 2.5Volts, Tamb=+25C (unless otherwise specified)
Symbol Iout Parameter Isink Short-circuit Output current coming in the op-amp. See fig-17 for more details Isource Output current coming out from the opamp. See fig-18 for more details Test Condition Output to GND Tmin. < Tamb < Tmax. Output to GND Tmin. < Tamb < Tmax. -340 Min. 360 Typ. 453 427 -400 -350 mA Max. Unit
Noise and distortion
eN Equivalent Input Noise Voltage see application note on page 13 Equivalent Input Noise Current (+) see application note on page 13 Equivalent Input Noise Current (-) see application note on page 13 Spurious Free Dynamic Range The highest harmonic of the output spectrum when injecting a filtered sine wave F = 100kHz F = 100kHz F = 100kHz AV = +2, Vout = 2Vp-p, RL = 100 F = 10MHz F = 20MHz F = 100MHz F = 150MHz 1.3 22 16 nV/Hz pA/Hz pA/Hz
iN
S FD R
-78 -73 -48 -37
dBc
Table 4. Closed-loop gain and feedback components
VCC (V) Gain +10 -10 +2 2.5 -2 +1 -1 270 300 260 530 1500 600 180 38 280 Rfb () 200 200 300 -3dB Bw (MHz) 280 270 1000 0.1dB Bw (MHz) 50 45 160
4/19
Electrical Characteristics
Figure 1. Frequency response, positive gain
24 22 20 18 16 14 12 10 8 6 4 2 0 -2 -4 S ma ll Signal -6 V cc =5 V -8 L oad= 100 - 10 1M
T SH 330
Figure 4. Frequency response, negative gain
24 22 20 18 16 14 12 10 8 6 4 2 0 -2 -4 Sm all Signal -6 Vc c= 5V -8 Loa d= 100 -10 1M
G ain= +10
Gain =-10
G ain=+ 4
G ain= -4
G ain (dB)
G ain= +2
Gain (dB)
Gain= -2
Ga in=+ 1
G ain= -1
10 M
1 00M
1G
1 0M
10 0M
1G
Fr eq uen cy (Hz)
F requ ency (Hz)
Figure 2. Gain flatness, gain=+4
12,2
Figure 5. Gain flatness, gain=+2
6,2
12,0
Gain Flatness (dB)
Gai n Flatness (dB)
6,0
11,8
Vin
+ -
Vout
Vi n
5,8
8k2 1pF
+ -
Vout
11,6
8k2 22pF
300R 100R
300R 300R
5,6
11,4
G ain=+ 4, Vcc=5V, Small Signal
Gain=+2, Vcc=5V, Small Signal
1M
1 0M
100 M
5,4 1M
10M
100M
1G
Fr equency (Hz)
Frequency (Hz)
Figure 3. Compensation, gain=+2
10 8 6 4 2 0 -2 -4 -6 -8 -10 -12 -14 -16 -18 -20 -22 1M
Gain=+2, Vcc=5V, Small Signal
8k2 1pF Vi n
+ -
Figure 6. Compensation, gain=+4
16 14 12 10 8 6 4 2 0 -2 -4 -6 -8 -10 -12 -14 -16 1M
8k2 22pF 3 0 0R 1 0 0R V in
+ -
G ain (dB)
G ain (dB)
Vout
Vout
300R 300R
G ain= + 4, Vcc=5V, Small Signal
10 M 100M 1G
10 M
100M
1G
Fr equency (Hz)
Fr equency (Hz)
5/19
TSH330
Figure 7. Compensation, gain=+10
24 22 20 18 16 14
Electrical Characteristics
Figure 10. Quiescent current vs. Vcc
20 15 10 5
Icc(+)
Gai n (dB)
12 10 8 6 4 2 0 -2 -4 -6 -8 1M
Gain=+10, Vcc=5V, Small Signal 15pF Vin
+ -
Vout
Icc (mA)
0 -5 - 10 - 15 - 20 - 25
200R 22R
G a in = + 2 Vc c = 5V Inpu t to ground, no load
1 ,50 1,7 5
Icc(-)
10 M
100M
1G
- 30 1,2 5
2,0 0
2 ,25
2,5 0
Fr eq uen cy (Hz)
+/-Vcc (V)
Figure 8. Input current noise vs. frequency
150 140 130 120 110 100
Figure 11. Input voltage noise vs. frequency
4.0
3.5
Neg. Current Noise
3.0
in (pA/VHz)
80 70 60 50 40 30 20 10 1k
Pos. Current Noise
en (nV/VHz)
100k 1M 10M
90
2.5
2.0
1.5
10k
1.0 1k
10k
100k
1M
10M
Frequency (Hz)
Frequency (Hz)
Figure 9. Output amplitude vs. load
4,0
Figure 12. Noise figure
40 35 30 25
3,5
Vout max. (Vp-p)
NF (dB)
Fr e q = ? Gain=+ 2 Vcc= 5V
20 15 10
3,0
2,5
5 0 1 10 100 1k 10k
V cc= 5 V
100k
2,0 10 100 1k 10k 100k
Load (ohms)
Rsource (ohms)
6/19
Electrical Characteristics
Figure 13. Output amplitude vs. frequency
5
T SH 330
Figure 16. Distortion vs. amplitude
-20 -25 -30 -35 -40
4
HD2 & HD3 (dBc)
V out max. (Vp-p)
-45 -50 -55 -60 -65 -70 -75 -80 -85 -90 -95 -100
3
HD2
2
1
0 1M
G a in = + 2 Vcc=5V Load = 100
10 M 100M 1G
HD3
0 1 2 3
G a in = + 2 Vc c =5 V F = 3 0 MH z L oad= 100
4
Fr equency (Hz)
Output Amplitude (Vp-p)
Figure 14. Distortion vs. amplitude
-20 -25 -30 -35 -40
Figure 17. Isink
600 550 500 450 400
RG +2 .5 V VO L
withou t load
+
- 1V
_
- 2.5 V
Isink V
HD2 & HD3 (dBc)
-45
Isink (mA)
-50 -55 -60 -65 -70 -75 -80 -85 -90 -95 -100 0 1 2 3 4
350 300 250 200
Amp lifier in open loop withou t load
HD2
HD3
G a in = + 2 Vc c=5 V F=1 0M Hz Loa d= 10 0
150 100 50 0 - 2,0 - 1,5 - 1,0 - 0,5 0,0
Output Amplitude (Vp-p)
V (V)
Figure 15. Distortion vs. amplitude
-20 -25 -30 -35 -40
Figure 18. Isource
0 -50 -100 -150
HD2 & HD3 (dBc)
-50 -55 -60 -65 -70 -75 -80 -85 -90 -95 -100 0 1 2 3 4
Isource (mA)
-45
-200 -250 -300 -350 -400 -450 -500 -550 -600 0,0 0,5 1,0 1 ,5 2,0
RG +1V +2.5 V VO H
HD2
+
_
- 2.5 V
without loa d
Iso urc e V
HD3
Gain =+2 Vc c=5V F =2 0MH z Lo ad= 100
Amp lifier in open loop with ou t loa d
Output Amplitude (Vp-p)
V (V)
7/19
TSH330
Figure 19. Slew rate
Electrical Characteristics
Figure 22. CMR vs. temperature
2,0
64 62
O utput Response (V)
1,5
60 58
CMR (dB)
Gain=+2 Vc c=5V Load=100
-1ns 0s 1ns 2ns 3ns
1,0
56 54 52 50
0,5
0,0 - 2ns
48 46
Gain=+1 Vcc=5V Load=100
-4 0 -20 0 20 40 60 80 100 120
Time (ns)
Tem perature (C)
Figure 20. Reverse isolation vs. frequency
0
Figure 23. SVR vs. temperature
85
80 -20 75
Gai n (dB)
-40
SVR (dB)
70
-60
65
60 -80
- 100 1M
Sm all Signal Vc c=5V Loa d= 100
1 0M 10 0M 1G
55
G a in = + 1 Vcc=5 V Loa d= 10 0
-4 0 -20 0 20 40 60 80 1 00 120
50
F requ ency (Hz)
Tem perature (C)
Figure 21. Bandwidth vs. temperature
1,3 1,2
Figure 24. ROL vs. temperature
200
180 1,1 1,0 0,9 0,8 0,7 0,6 0,5 -4 0 - 20 0 20 40 60 80 10 0 1 20
Bw (GHz)
ROL (M)
G a in = + 2 Vcc =5V Load =1 00
160
140
120
Open Loop Vcc=5V
100 -40 - 20 0 20 40 60 80 100 120
Temperature (C)
Tem peratur e (C)
8/19
Electrical Characteristics
Figure 25. I-bias vs. temperature Figure 28. Icc vs. temperature
20 24 15 22 20 18
Ib(+)
Icc(+)
T SH 330
10 5 0
IBIAS (A)
16 14 12 10 8 6
Ib(-)
ICC (mA)
-5 -10
Icc(-)
-15 -20
G ai n= + 1 V c c = 5V L oa d= 100
-4 0 - 20 0 20 40 60 80 10 0 12 0
Gain=+1 Vcc=5V no Load -30 In+/In- to GND
-25 -35 -40 -20 0 20 40 60 80 10 0 120
Temperature (C)
Temperature (C)
Figure 26. Vio vs. temperature
100 0
Figure 29. Iout vs. temperature
600
400 80 0 200
Isource
VIO (micro V)
Iout (mA)
60 0
0
40 0
- 200
Isink
- 400 20 0
Open Loop Vcc=5V Load =10 0
-40 - 20 0 20 40 60 80 1 00 12 0
- 600
0
Output: short-circuit Gain=+1 Vcc=5V
-4 0 -20 0 20 40 60 80 100 120
- 800
Temper atur e (C)
Temperature (C)
Figure 27. VOH & VOL vs. temperature
2
1
V OH
VOH & OL (V)
0
-1
VO L
-2
-3
Gain=+1 Vcc=5V Load=100
-20 0 20 40 60 80
-4 -40
Tem peratur e (C)
9/19
TSH330
Evaluation Boards
3 Evaluation Boards
An evaluation board kit optimized for high-speed operational amplifiers is available (order code: KITHSEVAL/STDL). The kit includes the following evaluation boards, as well as a CD-ROM containing datasheets, articles, application notes and a user manual:
SOT23_SINGLE_HF
BOARD: Board for the evaluation of a single high-speed op-amp in SOT23-5
package.
SO8_SINGLE_HF: SO8_DUAL_HF: SO8_S_MULTI:
Board for the evaluation of a single high-speed op-amp in SO8 package.
Board for the evaluation of a dual high-speed op-amp in SO8 package.
Board for the evaluation of a single high-speed op-amp in SO8 package in inverting and non-inverting configuration, dual and single supply. Board for the evaluation of a triple high-speed op-amp in SO14 package with video application considerations.
SO14_TRIPLE:
Board material:
2
layers (r=4.6) 1.6mm thickness: 35m
FR4
epoxy
copper
Figure 30. Evaluation kit for high-speed op-amps
10/19
Power Supply Considerations
T SH 330
4 Power Supply Considerations
Correct power supply bypassing is very important for optimizing performance in high-frequency ranges. Bypass capacitors should be placed as close as possible to the IC pins to improve high-frequency bypassing. A capacitor greater than 1F is necessary to minimize the distortion. For better quality bypassing, a capacitor of 10nF can be added using the same implementation conditions. Bypass capacitors must be incorporated for both the negative and the positive supply. For example, on the SO8_SINGLE_HF board, these capacitors are C6, C7, C8, C9. Figure 31. Circuit for power supply bypassing
+VCC + 10nF
10microF
+ 10nF
10microF + -VCC
Single power supply
In the event that a single supply system is used, new biasing is necessary to assume a positive output dynamic range between 0V and +VCC supply rails. Considering the values of VOH and VOL, the amplifier will provide an output dynamic from +0.9V to +4.1V on 100 load. The amplifier must be biased with a mid-supply (nominally +VCC/2), in order to maintain the DC component of the signal at this value. Several options are possible to provide this bias supply, such as a virtual ground using an operational amplifier or a two-resistance divider (which is the cheapest solution). A high resistance value is required to limit the current consumption. On the other hand, the current must be high enough to bias the non-inverting input of the amplifier. If we consider this bias current (55A max.) as the 1% of the current through the resistance divider to keep a stable mid-supply, two resistances of 470 can be used. The input provides a high pass filter with a break frequency below 10Hz which is necessary to remove the original 0 volt DC component of the input signal, and to fix it at +VCC/2.
Figure 32 illustrates a 5V single power supply configuration for the SO8_SINGLE evaluation board (see Evaluation Boards on page 10).
11/19
TSH330
Power Supply Considerations
A capacitor CG is added in the gain network to ensure a unity gain in low-frequency to keep the right DC component at the output. CG contributes to a high-pass filter with Rfb//RG and its value is calculated with a consideration of the cut off frequency of this low-pass filter. Figure 32. Circuit for +5V single supply
+5V 10F IN +5V R1 470 Rfb R2 470 + 1F 10nF + RG CG Rin 1k
+
_
100F
OUT 100
12/19
Noise Measurements
T SH 330
5 Noise Measurements
The noise model is shown in Figure 33, where:
eN: iNn: iNp:
input voltage noise of the amplifier negative input current noise of the amplifier positive input current noise of the amplifier
Figure 33. Noise model
+
R3
iN+
_
output HP3577 Input noise: 8nV/Hz
N3
iN-
eN
N2
R1
R2
N1
The thermal noise of a resistance R is:
4kTRF
where F is the specified bandwidth. On a 1Hz bandwidth the thermal noise is reduced to
4kTR
where k is the Boltzmann's constant, equal to 1,374.10-23J/K. T is the temperature (K). The output noise eNo is calculated using the Superposition Theorem. However eNo is not the simple sum of all noise sources, but rather the square root of the sum of the square of each noise source, as shown in Equation 1:
eNo = 2 2 2 2 2 2 V1 + V2 + V3 + V 4 + V5 + V6
Equation 1
eN o
2
2 2 2 2 2 2 2 R 22 R -- 2 = e N × g + i N n × R 2 + i N p × R 3 × g + ------- × 4 k T R 1 + 4 k T R 2 + 1 + -----2 × 4 k T R 3 R1 R1
Equation 2
13/19
TSH330
Noise Measurements
The input noise of the instrumentation must be extracted from the measured noise value. The real output noise value of the driver is:
eNo = 2 2 ( Me a s u r e d ) (i n s t r u m e n t a t i o n )
Equation 3
The input noise is called the Equivalent Input Noise as it is not directly measured but is evaluated from the measurement of the output divided by the closed loop gain (eNo/g). After simplification of the fourth and the fifth term of Equation 2 we obtain:
eNo 2 2 2 2 2 2 2 2 R22 = e N × g + i N n × R 2 + i N p × R 3 × g + g × 4 k T R 2 + 1 + ------- × 4 k T R 3 R1
Equation 4
Measurement of the Input Voltage Noise eN
If we assume a short-circuit on the non-inverting input (R3=0), from Equation 4 we can derive:
eNo = 2 2 2 2 eN × g + iNn × R2 + g × 4kTR2
Equation 5
In order to easily extract the value of eN, the resistance R2 will be chosen to be as low as possible. In the other hand, the gain must be large enough:
R3=0, gain: g=100
Measurement of the Negative Input Current Noise iNn
To measure the negative input current noise iNn, we set R3=0 and use Equation 5. This time the gain must be lower in order to decrease the thermal noise contribution:
R3=0, gain: g=10
Measurement of the Positive Input Current Noise iNp
To extract iNp from Equation 3, a resistance R3 is connected to the non-inverting input. The value of R3 must be chosen in order to keep its thermal noise contribution as low as possible against the iNp contribution:
R3=100W, gain: g=10
14/19
Intermodulation Distortion Product
T SH 3 3 0
6 Intermodulation Distortion Product
The non-ideal output of the amplifier can be described by the following series:
2 n V o u t = C 0 + C 1 Vi n + C 2 V i n + ... Cn V i n
due to non-linearity in the input-output amplitude transfer, where the input is Vin=Asint, C0 is the DC component, C1(Vin) is the fundamental and Cn is the amplitude of the harmonics of the output signal Vout. A one-frequency (one-tone) input signal contributes to harmonic distortion. A two-tone input signal contributes to harmonic distortion and to the intermodulation product. The study of the intermodulation and distortion for a two-tone input signal is the first step in characterizing the driving capability of multi-tone input signals. In this case:
Vi n = A sin 1 t + A sin 2 t
t hen:
2 n V o u t = C 0 + C1 ( A sin 1 t + A sin 2 t ) + C 2 ( A sin 1 t + A sin 2 t ) ... + C n ( A sin 1 t + A sin 2 t )
From this expression, we can extract the distortion terms, and the intermodulation terms form a single sine wave: second-order intermodulation terms IM2 by the frequencies (1-2) and (1+2) with an amplitude of C2A2 and third-order intermodulation terms IM3 by the frequencies (21-2), (21+2), (- 1+22) and (1+22) with an amplitude of (3/4)C3A3. The measurement of the intermodulation product of the driver is achieved by using the driver as a mixer by a summing amplifier configuration (see Figure 34). In this way, the non-linearity problem of an external mixing device is avoided. Figure 34. Inverting summing amplifier (using evaluation board SO8_S_MULTI)
Vin1 Vin2
R1
R fb
R2
_
Vout
+
100
R
15/19
TSH330
The Bias of an Inverting Amplifier
7 The Bias of an Inverting Amplifier
A resistance is necessary to achieve a good input biasing, such as resistance R shown in Figure 35. The magnitude of this resistance is calculated by assuming the negative and positive input bias current. The aim is to compensate for the offset bias current, which could affect the input offset voltage and the output DC component. Assuming Ib-, Ib+, Rin, Rfb and a zero volt output, the resistance R will be:
Rin × Rfb R = ---------------------R i n + Rf b
Figure 35. Compensation of the input bias current
R fb
Ib-
Rin
_
Vcc+ Output
+
Ib+ R Vcc-
Load
16/19
Active Filtering
T SH 330
8 Active Filtering
Figure 36. Low-pass active filtering, Sallen-Key
C1
R1 IN
R2 C2
+
OUT
_
100
Rf b RG
From the resistors Rfb and RG we can directly calculate the gain of the filter in a classical non-inverting amplification configuration:
Rf b A V = g = 1 + --------Rg
We assume the following expression as the response of the system:
V o u tj g T j = ------------------- = --------------------------------------------V i nj 2 j ( j) 1 + 2 ------ + ------------ c 2 c
The cut-off frequency is not gain-dependent and so becomes:
1 ---c = -------------R-------------------2 R1 2C 1C
The damping factor is calculated by the following expression:
1 = -- c ( C 1 R1 + C 1 R2 + C 2 R 1 C 1 R 1 g ) 2
The higher the gain, the more sensitive the damping factor is. When the gain is higher than 1, it is preferable to use some very stable resistor and capacitor values. In the case of R1=R2=R:
Rfb 2 C 2 C1 --------Rg = ----------------------------------2 C1 C2
Due to a limited selection of values of capacitors in comparison with resistors, we can fix C1=C2=C, so t hat :
Rfb 2 R 2 R1 --------Rg = ----------------------------------2 R1 R2
17/19
TSH330
Package Mechanical Data
9 Package Mechanical Data
SO-8 MECHANICAL DATA
DIM. A A1 A2 B C D E e H h L k ddd 0.1 5.80 0.25 0.40 mm. MIN. 1.35 0.10 1.10 0.33 0.19 4.80 3.80 1.27 6.20 0.50 1.27 0.228 0.010 0.016 TYP MAX. 1.75 0.25 1.65 0.51 0.25 5.00 4.00 MIN. 0.053 0.04 0.043 0.013 0.007 0.189 0.150 0.050 0.244 0.020 0.050 inch TYP. MAX. 0.069 0.010 0.065 0.020 0.010 0.197 0.157
8 (max.)
0.04
00160 23/C
18/19
Revision History
T SH 330
10 Revision History
Date Revision Description of Changes
Oct. 2004 Dec. 2004 June 2005
1 2 3
First release corresponding to Preliminary Data version of datasheet. Release of mature product datasheet.
Table 1 on page 2 - Rthjc: Thermal Resistance Junction to Ambient replaced by Thermal Resistance Junction to Case
Information furnished is believed to be accurate and reliable. However, STMicroelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of STMicroelectronics. Specifications mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied. STMicroelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of STMicroelectronics. The ST logo is a registered trademark of STMicroelectronics All other names are the property of their respective owners 2004 STMicroelectronics - All rights reserved STMicroelectronics group of companies Australia - Belgium - Brazil - Canada - China - Czech Republic - Finland - France - Germany - Hong Kong - India - Israel - Italy - Japan Malaysia - Malta - Morocco - Singapore - Spain - Sweden - Switzerland - United Kingdom - United States of America www.st.com
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